Since the early days of radio, system designers of radiating systems have been encumbered by antenna size restrictions at operating frequencies below 30 MHz. Resonant antennas, which have a resistive input impedance, are prohibitively large, while smaller antennas are highly reactive and inhibit the transfer of power to radiation energy, especially when a wide bandwidth is required. The terms large and small are used to refer to antenna size relative to wavelength. It has been a long held belief of RF engineers that the bandwidth for efficient energy transfer from the source to a highly reactive load is inversely proportional to the Q of that load. That statement is true when the interface or matching network is restricted to passive components. As will be described, the present invention overcomes that limitation by use of active components to deliver broadband RF power to the real part of a highly reactive load. Although a specific application is the excitation of small antennas, the invention can be useful in other applications where it is desired to provide more effective energy coupling to a high-Q load of any type.
The equivalent circuits of common small antennas are shown in FIG. 1a, which represents a loop antenna, and FIG. 1b, which represents a dipole or whip-type monopole antenna. Antenna designers typically model an antenna with a fictitious resistor called the radiation resistance. Since the only element that can absorb or dissipate power is a resistive element the radiated power, or that absorbed by space, is modeled by the radiation resistance. The value of the radiation resistance and its associated reactance depends on the geometry of the antenna and its dimensions with respect to a wavelength. The ratio of reactance to resistance or Q is given by equation (1). ##EQU1## where .lambda. is the wavelength and L is the physical size of the antenna. For example, a 10 foot antenna operating at 2 MHz has a Q of about 20,000.
Power transfer between a source and a high Q load is difficult because of voltage division between the load reactance and the small load resistance. In the above example, a power amplifier that provided an output at 20,000 volts would only produce 1 volt across the radiation resistance and result in a very small amount of radiated power.
A summary of possible approaches using existing technology is shown in FIGS. 2a and 2b. When high efficiency is desired, the narrowband approach of FIG. 2a can be used. High efficiency is achieved by use of a high-efficiency non-linear power amplifier A feeding a tuner T, which tunes the antenna by using the opposite type reactor in series with the antenna reactance to yield a net reactance of zero at resonance. The reactances can only cancel at one frequency, and the 1/2 power bandwidth, which occurs when the circuit's net reactance is equal to that of the resistance, is equal to 1/Q. With this approach, it is possible to use a non-linear, highly efficient (e.g., class C) amplifier because all higher harmonics caused by the non-linear distortion are filtered by the tuned circuit. In theory, if there were no losses associated with the tuning reactance, it would be possible to achieve 100% efficiency of power transfer at one frequency using this approach. In the aforementioned example at 2 MHz, the 1/2 power bandwidth would be only 100 Hertz.
When, using prior technology, it is desirable to radiate power over a wide instantaneous frequency bandwidth, the approach indicated in FIG. 2b can be used. This approach uses a linear power amplifier A with no tuning reactor. The amplifier could be either class A or the more efficient push-pull class B or class AB. The amplifier must be linear because no filtering of harmonics caused by distortion occurs in this circuit. The efficiency of this approach, using a class B push-pull amplifier, is approximately equal to 1/Q. Given the example of a 10-foot antenna at 2 MHz, a 1000 watt class B push-pull amplifier could deliver over a broad frequency range only 50 milliwatts to radiation.
Efficiency and bandwidth can be traded off using an intermediate approach. That is accomplished by resistive loading (placing a larger resistor in series with the radiation resistance.) The bandwidth is increased because the effective Q of the load is lowered, however, most of the power delivered to the load is dissipated in the series resistor, thereby decreasing efficiency. As a practical matter, almost all circuits have resistive loading which results from either loss in the tuning reactor, and/or, conductor losses in the antenna. If a tuned circuit Q is defined as the reactance divided by all loss resistance, then efficiency and bandwidth can be defined by equations (2) and (3) ##EQU2## where Q.sub.c is the tuned circuit Q. It is assumed that Q.sub.c &lt;&lt;Q.sub.A. In a practical antenna, the highest achievable circuit Q at 2 MHz might be about 500. In the previous example, the bandwidth would be 4 KHz, while the efficiency would be 0.025 (1000 W amplifier delivers 25 W radiation).
In the case of the broadband linear amplifier, most of the power delivered by the DC power source is dissipated by the amplifier output device. Instantaneous power dissipation by any device is defined as the voltage across the device multiplied by the current flowing through it at the same instant in time. The load line of each device in a class B push-pull amplifier is shown in FIG. 3, wherein I represents the device output current and V represents the device voltage. As shown, the CP lines represent control parameter characteristics of the amplifier, V.sub.Q is the bias voltage, V.sub.p is the peak voltage across the device, I.sub.p is the peak device current and L is the load line. The circular load line L occurs when the amplifier drives a reactive load (voltage and current waveforms in quadrature). Throughout most of the cycle, power is dissipated because both voltage and current occur simultaneously in the device. Average dissipation can be computed through integration of the instantaneous power dissipation over 1/2 cycle. The result of that integration is: ##EQU3## where V.sub.Q is the quiescent bias voltage and X.sub.L is the reactance of the load.
If A is the peak amplitude of the sinusoidal voltage waveform, then V.sub.Q =A is the optimum bias, considering power dissipation. ##EQU4## The output power delivered to the real part of the load is found by voltage division between the load reactance and resistance. ##EQU5## Efficiency is approximately equal to the power delivered to the load divided by that dissipated in the source. ##EQU6##
It can be shown that efficiency in this antenna context is limited by conservation of energy. Amplifier output devices that operate in a switch mode (ON: High Current, Voltage=0; OFF: High Voltage, Current=0) theoretically dissipate no power because the instantaneous multiplication of voltage and current is always zero. Class D and E amplifiers use this principle to achieve high efficiency. Such amplifiers are intended for use with resistive loads and use intermediate filter networks in order to eliminate the harmonics that are generated by the switch. The filter network transforms the square-wave type switch output to the desired sinusoidal output waveform.
Attempts have been made in the past to improve the efficiency of the linear amp/reactive antenna combination by using switchmode output devices. As opposed to the class D and class E amplifiers where the switch frequency is equal to the output frequency, a switchmode arrangement may use a switch frequency that is faster than the desired output frequency. A low pass filter can be placed between the amplifier and load. The filter, in effect, averages the switch output voltage. Therefore, it is possible to vary the filter output voltage by adjusting the ratio of ON to OFF times of the switch. This approach is wideband up to a reasonable percentage of the low pass filter cutoff. A possible wideband amplifier/small antenna approach could use this technique in order to synthesize the high voltage at the input terminals of a reactive antenna. However, it has been found that only insignificant efficiency improvements are possible. One explanation addresses the effect of the low pass filter input impedance when driving a reactive load. The input impedance is always reactive at the switch mode frequency. Therefore, the switch current and voltage must be in quadrature (instead of anti-phase) and instantaneous power is always dissipated.
All known attempts at modifying such a filter in order to improve efficiency have been unsuccessful. The result can be attributed to the fundamental principle of conservation of energy. FIG. 4 shows the basic circuit and its waveforms. As shown, a small antenna is connected across the output terminals of a power source P. Since the source output voltage is much larger than V.sub.o across the resistive part of the load L, nearly all the input voltage, or current, is across or through the reactance. Any reactor is an energy storage device with the following amounts of energy stored: EQU E=1/2 CV.sup.2 (capacitor) E=1/2LI.sup.2 (inductor) (9)
As the waveform cycles through its periodic variation, there must be a large peak of stored energy in the reactor. When the waveform cycles back through zero, the stored energy will be equal to zero. Conservation of energy mandates that the peak stored energy cannot disappear. It must either be transferred to another storage media, or dissipated. In the case of the broadband amplifier, where there is no other reactor for storage,it must be dissipated by the power source. In the case of a narrowband, tuned amplifier, energy is transferred to the tuning reactor, which is of the opposite type of reactance from the antenna reactor. When one reactance has peak energy, the other has zero and at intermediate times, the energy is distributed between the two reactors. At all times, the total circuit energy is equal to the peak value, as computed by the peak voltage or current that must be imposed on the antenna load. No energy is dissipated in this ideal case. It is transferred back and forth at the resonant frequency of the tuned circuit. Therefore, this approach is very efficient, but provides only very narrowband operation.
So far as is known, no prior broadband approach has effectively incorporated another reactor for storage of transferred energy. Therefore, the output devices have dissipated the peak energy at a rate of twice each cycle. A limit on the efficiency of a broadband amplifier can be computed from energy considerations. Since power is the rate of energy dissipation, average power can be found by integrating the rate of energy dissipation over a cycle of the output waveform. The result of that computation, where P.sub.D is the dissipated power, is: ##EQU7## A comparison with the class B push-pull amplifier shows that this type of circuit can theoretically perform within 25% (6 dB) of the maximum achievable efficiency.
It is, therefore, an object of this invention to provide radiating systems able to achieve both high efficiency and broad-band operation in conjunction with a high Q circuit and, more particularly, in combination with a small antenna.
It is a further object to provide new forms of synthesizer radiating systems and, more particularly, such systems having a transmit antenna which is interactive with a waveform synthesizer amplifier providing active control of one or more of the direction, rate and level of energy transferred between a radiating element with a reactive characteristic and an energy storage element with either an opposite or a similar reactive characteristic.
Additional objectives are to provide new and improved radiating systems which avoid one or more limitations of prior systems, as well as systems which achieve wider bandwidth operation or improved efficiency, or both, in the coupling of energy to a high-Q load, which may be a radiating element.